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AQ: FETs in ZVS bridge

Had run into a very serious field failure issue a decade ago due to IXYS FETs used in a phase-shifted ZVS bridge topology. Eventually, the problem was tracked to failure of the FETs’ body diode when the unit operated at higher ambient temperature.

When FETs were first introduced for use in hard switching applications, it was quickly discovered that under high di/dt commutating conditions, the parasitic bipolar transistor that forms the body diode can turn on resulting in catastrophic failure (shorting) of the FET. I had run into this issue in the mid ’80s and if memory serves me correctly, IR was a leader in making their FET body diodes much more robust and capable of hard commutation. Having had this experience with FET commutation failures and after exhausting other lines of investigation which showed no problem with the operation of the ZVS bridge, I built a tester which could establish an adjustable current through the body diode of the FET under test followed by hard commutation of the body diode.

Room temperature testing of the suspect FET showed the body diode recovery characteristic similar to that of what turned out to be a more robust IR FET. Some difference was seen in the diode recovery as the IXYS FET was a bit slower and did show higher recovered charge. However, was unable to induce a failure in either the IXYS or IR FET even when commutating high values of forward diode current up to 20A when testing at room temperature.

The testing was then repeated in a heated condition. This proved to be very informative. The IXYS FETs were found to fail repeatedly with a case temperature around 80C and forward diode current prior to commutation as low as 5A. In contrast, the IR devices were operated to 125C case temp with forward diode currents of 10A without failure.

This confirmed a high temperature operating problem of the IXYS FETs associated with the body diode. Changing to the more robust IR devices solved the field failure issue.
Beware when a FET datasheet does not provide body diode di/dt limits at elevated ambient.

A more complete explanation of the FET body diode failure mechanism in ZVS applications can be found in application note APT9804 published by Advanced Power Technology.

I believe FETs can be reliably used in ZVS applications if the devices are carefully selected and shown to have robust body diode commutation characteristics.

AQ: Heavily discontinuous mode flyback design

With a heavily discontinuous mode flyback design, the transformer’s ac portion of current can be larger than the dc portion. When a high perm material is used for the transformer core, the required gap can be quite large in order to reach the low composite permeability required while the core size will likely be driven by winding and core loss considerations rather than just simply avoiding saturation. Normally the gap is put in the center leg only (with E type topology cores) in order to minimize the generation of stray fields. However, in designs such as yours (high ac with a high perm core) the needed core gap can lead to a relatively large fringing zone through which foil or solid wire may not pass without incurring excessive, unacceptable loss. Possible solutions are to use Litz wire windings or inert spacers (e.g., tape) around the center leg in order to keep the windings far enough away from the gap (the rule of thumb is 3 to 5 gap lengths, which can eat up a lot of the window area).

It is mainly for these reasons that placing half the gap in an E type core’s outer legs might be worth the trouble of dealing with the magnetic potential between the core halves (and you have seen first hand what trouble an ill designed shield band can be).

To avoid eddy current losses, the shield band should be spaced well away from the outer leg gap, probably 5 gap lengths or more. Also to be a really effective magnetic shield, it should be 3 to 5 gap lengths thick.

Bear in mind that with a high frequency, high ac current inductor design proximity effects in the winding may become very significant. This is why many of these type of inductors have single layer windings or winding wound with Litz wire (foil is the worst winding type here). One advantage of an equally gapped E type core design is that the proximity effect on the windings is significantly less because there are two gaps in series (a quasi distributed gapped core design). Not only layer-to-layer, but turn-to-turn proximity effects can sometimes be problematic in an ac inductor (or flyback) design. Just as with the gap, these are reduced by adding appropriate spacing, for example making the winding coil loose or winding it bifilar with a non-conductive filament.

AQ: Power converter trend

The trend toward lower losses in power converters is not apparent in all of the applications of power converters. It is also not apparent that the power converter solution and its losses for a given market will be the same when it comes to losses. In terms of the market shift that you mention, Prof. the answer is probably that each market is becoming split into a lower efficiency and higher efficiency solution.

From my limited view the reason for this is the effort and time required to do the low loss development. The early developers of low loss converters are now ahead and those that were slower may never catch them. This gap is in a number of converter markets widening, with both higher loss and lower loss offerings continuing to be used and sold. This split is not apparent with different levels of development or geographically.

Some markets already have very efficient solutions, other markets not so efficient and others had high power loss solutions. The customers accepted these solutions. The path to lower loss converters is for some markets not yet clear and in some markets the requirement may never actually become real.

It does seem that there is a real case to make for any power converter market splitting in two as the opportunities presented by lowering the power loss are taken.

All low loss converters present significant challenges and are all somewhat esoteric.

For me power supply EMI control consists of designing filtering for differential and common mode conducted emissions. The differential mode filtering attenuates the primary side differential lower frequency switching current fundamental & harmonic frequencies. The common mode filtering provides a low impedance return path for high frequency noise currents resulting from high dV/dt transitions during switching transitions present on the power semiconductors (switching mosfet drain, rectifier cathods). These noise currents ring at high frequencies as they oscillate in the uncontrolled parasitic inductance and capacitance associated with their return to source path. Shortening and damping this return path allows the high frequency noise currents to return locally instead of via the measurement copper bench and conducted emi current or voltage (LISN) probe as well as providing a more damped ringing frequency. Shorting this return path has the added benefit of decreasing radiated emissions. In addition proper layout of the power train so as to minimize the loop area associated with both the primary and secondary side switching currents minimizes the associated radiated emissions.

When I mentioned the criticism of resonant mode converter as related to the challenges of emi filitering I was referring to the additional differential mode filtering required. For example if you take a square wave primary side current waveform and analyze the differential frequency content the fundamental magnitude with be lower and there will be higher frequency components as compared to a purely resonant approach at the same power level. It is normally the lower frequency content that has to be filtered differentially.

Given these differences the additional emi filtering volume/cost of the resonant approach may pose a disadvantage.

AQ: flyback & boost applications

For flyback & boost applications, powder cores such as Kool-mu, Xmu, etc… are usually best performing and lowest cost. Even these may need to be gapped and if CCM operation is required, a “stepped-gap” is preferred to allow a large load compliance. Center stepped gaps reduce the fringe flux greatly as there is never a complete gap, only localized saturation. This permits the inductor’s value to “swing” more and accommodate the required operation.
With only the center leg with a gap, the outer copper band can be applied without significant loss.

To explore further, dissimilar core materials can be used in parallel, ferrite & powdered types, such that different materials provide function at different operating points within the same construction. Some decades ago, we had some high power projects that utilized fixed magnets within a ferrite’s gap to provide a flux bias offset for a forward topology.

Abe Pressman wasn’t big on exploring magnetic losses, however he operated at lower frequencies than are typical today. MPPs are great with large DC bias, but suffer high loss if AC swing is large and fast. Toroids also have the least efficient winding window, however, they are best to mitigate emi.

AQ: Grain Storage system

A Grain Storage system usually consists of the following elements

1. A means of measuring Grain coming in and out- Usually a truck scale or a bulk weighing system. In addition some applications require measuring grain between transfers to different bins and a bulk weighing system is usually used for that.

2. A means of transferring between different operations or storage location- Conveyors, screws, buckets, pneumatic, wheel house.

3. Dust Collection

4. And Equipment for the operations that will be performed, drying, cleaning, screening, grading, sampling, roasting, steaming, packaging etc.

A system I have just completed was 24 containers + 3 buildings for storage with 76 conveyors, 3 drop-off and 3 loading points. Connection to ERP system and weigh scales to weigh trucks and send them to the correct bay. Local HMI on each bay ensured correct lorry goes to correct bay. Main conveyor runs are automatically selected. Manual option to run all conveyors to move grain around.

System used Ethernet infrastructure with hubs mounted strategically around tank farm. Also implemented soft starter with Ethernet connectivity, thus allowing easy monitoring of current consumption + for maintenance.

The future-proof design will allow customer to install level and humidity measurement in the future using the Analogue IO connected on Ethernet.

AQ: How to suppress chaotic operation in a DCM flyback at low load

I would like to share these tips with everybody.
A current mode controlled flyback converter always becomes unstable at low load due to the unavoidable leading edge current spike. It is not normally dangerous but as a design engineer I don’t like to look at it and listen to it.

Here are three useful and not patented tips.

First tip:
• Insert a low pass filter, say 1kohm + 100pF between current sense resistor and CS input in your control IC.
• Split the 1kohm in two resistors R1 to the fet and R2 to the control IC. R1 << R2.
• Insert 0,5 – 1pF between drain and the junction R1/R2. This can be made as a layer-to-layer capacitor in the PCB. It does not have to be a specific value.
• Adjust R1 until the spike in the junction in R1/R2 is cancelled.
You will see that the current spike is always proportional to the negative drain voltage step at turn-on. Once adjusted, the cancellation always follows the voltage step, and you some times achieve miracles with it. Cost = one resistor.

Second tip:
Having the low pass filter from first tip, add a small fraction of the gate driver output voltage to the current sense input, say 0,1V by inserting a large resistor from ‘Drive Out’ to ‘CS input’. The added low pass filtered step voltage will more or less conceal the current spike. You should reduce your current sense resistor accordingly. Cost = one resistor.

Third tip:
In a low power flyback, you some times just need an RC network or just an extra capacitor from drain to a DC point, either to reduce overshoot or to reduce noise. Connect the RC network or the capacitor to source, not to ground or Vcc. If you connect it to ground or Vcc, you will measure the added discharge current peak in the current sense resistor. Cost = nothing – just knowledge.

All tips can be used individually or combined => Less need for pre-load resistors on your output.

AQ: Determine coefficient of grounding

Determination of required grounding impedance is based on determination of coefficient of grounding which represents ratio of maximum phase voltage at phases which aren’t exposed by fault and line voltage of power network:

kuz=(1/(sqrt(3)))*max{|e(-j*2*π/3)+(1-z)/(2+z)|; |e(+j*2*π/3)+(1-z)/(2+z)|}
z=Z0e/Zde

where are:

kuz-coefficient of grounding,
z-ratio of equivalent zero sequence impedance viewed from angle of place of fault and equivalent direct sequence impedance viewed from angle of place of fault,
Z0e-equivalent zero sequence impedance viewed from angle of place of fault,
Zde-equivalent direct sequence impedance viewed from angle of place of fault.

So, after this explanation, you can get next conclusions:
if kuz=1 then power network is ungrounded because Z0e→∞, which is a consequence of existing more (auto) transformers with ungrounded neutral point than (auto) transformers with grounded neutral point (when kuz=1 then there aren’t (auto) transformers with grounded neutral point),
if kuz≤0,8 then power network is grounded because Z0e=Zde, which is a consequence of of existing more (auto) transformers with grounded neutral point than (auto) transformers with ungrounded neutral point.

Fault current in grounded power networks is higher than fault current in ungrounded power networks. By other side, in case of ungrounded power networks we have overvoltages at phases which aren’t exposed by fault, so insulation of this conductors could be seriously damaged or in best case it could become older in shorter time than it is provided by design what is the main reason for grounding of power networks.
Coefficient of grounding is very important in aspect of selecting of insulation of lighting arresters and breaking power of breakers, because of two next reasons:
1. in grounded power networks insulation level is lower than insulation level in ungrounded power networks,
2. in grounded power networks value of short circuit current is higher than value of short circuit current in ungrounded power networks.

AQ: How to design an Panel required for PLC / MCC / Drive

1. The regular industrial standard size panel available with most of the panel fabricator’s.
2. Type of protection (used to say as IP).
3. Spacing depends upon the Power handled by the conductors inside Panel and the ventilation system.
4. Cable Entry / Bus bar entry may depend on the application and site condition. it may be at rear/bottom or at the top.
5. When comes to Drive, if the site condition is too hot then an industrial ac is required usually attached at the side of the panel.
6. Drive to drive required spacing (Check the manual of the drive}, since the power switching activity take place inside the drive.
7. Provide required space for the transformers and AC-choke since they create magnetic flux in ac circuits.
8. Don’t mix the Control cable, Power cable, Signal cable and Communication cable together in the cable tray… Otherwise you will be wired…
9. Keep the control on mcb/mccb/mpcb in handy location. So that its easier for operator to control it frequently and not disturbing other circuits..
10. Plc will be acquiring the top position in the panel since there is nothing to do with it once installed. Just we will be monitoring the status.
11. Don’t place the Plc nearby to the incoming or outgoing heavy power terminals..
12. Mcc panel are easy thing to do, but do the exact calculation for the ACB selection in the incomer side. Since each feeder will be designed with tolerance level.

There being a lot more than 12 guidelines to follow. What
about back-up power for the PLC? What about internal heat flow considerations
(not just does it need an AC or not)? How much space between terminal blocks
and wireway? What about separate AC and instrument grounds? What about wireway
fill? What about wire labels? What about TSP shields? What about surge
protection? In my experience, there are plenty of people that can design a
panel but if they haven’t gone to the field with it then they haven’t been able
to learn from their design mistakes.

The best thing you can do is start your design but you
really need to be guided by an experienced designer.

AQ: Right Half Plane Pole

Very few know about the Right Half Plane Pole (not a RHP-Zero) at high duty cycle in a DCM buck with current mode control. Maybe because it is not really a problem.
It is said that this instability starts above 2/3 duty cycle – I think that must be with a resistive load. If loaded with a pure current source, it starts above 50% duty cycle.

Here is a little down-to-earth explanation:
If you run a buck converter at high duty cycle but DCM, it probably works fine and is completely stable. Then imagine you suddenly open the feedback loop, leaving the peak current constant and unchanged. The duty cycle will then rush either back to 50% or to 100% if possible. You now have a system with a negative output resistance – if Voltage goes up, the output current will increase.

You can see it by drawing some triangles on a piece of paper: A steady state DCM current triangle with an up-slope longer than the down-slope and a fixed peak value. Now, if you imagine that the output voltage rises, you can draw a new triangle with the same peak current. The up-slope will be longer, the down-slope will be shorter but the sum of times will be longer than in the steady state case. The new triangle therefore has a larger area than the steady state triangle, which means a higher average output current. So higher output voltage generates higher output current if peak current is constant. Loaded with a current source, it is clear that this is an unstable system, like a flipflop, and it starts becoming unstable above 50% duty cycle.

However, when you close the feedback loop, the system is (conditionally) stable and the loop gain is normally so high at the RHP Pole frequency that it requires a huge gain reduction to make it unstable.

It’s like when you drive on your bike. A bike has two wheels and therefore can tilt to either side – it is a system with a low frequency RHPP like a flipflop. If you stand still, it will certainly tilt to the left or to the right because you have no way to adjust your balance back. But if you drive, you have a system with feedback where you can immediately correct imbalance by turning the handlebars. As we know, this system is stable unless you have drunk a lot of beers.

AQ: What causes VFD driven motor bearing current?

There are several things involved, all with varying degrees of impact.

Large machines are – generally speaking – made of pieces (segments) because the circle for the stator and/or rotor core is too large to manufacture from a single sheet. This leads to some breaks in the magnetic flux path symmetry, both in the radial (right angles to the shaft) and axial (parallel to the shaft) directions.

For the most part, the windings of large machines are formed and installed by hand. This too can lead to symmetry issues, as the current paths are not identical which in turn will create some differences in the magnetic field flux.

Output waveforms from power electronics are only approximations of true sinusoids. The presence of additional harmonics distorts the sinusoidal nature and results in changes that are not symmetric in the magnetic field strength … which in turn means a non-symmetric flux distribution.

Two other items contribute to potentially damaging bearing currents as well. One of these is the Common Mode Voltage which is present (to some degree) in all drives. Essentially this is a signal that is present at both the drive input and output … I tend to think of it as an offset. It’s not something that traditional grounding addresses, and can create an elevated potential in the shaft which then discharges through the bearing path.

A second item is not related to the presence (or absence) of drives at all; it is related to the mechanical arrangement of the process drive train. For example, a shaft that has a sliding seal (like the felt curtain on a dryer section), or one that turns a blade against a gas or liquid (like a compressor) can generate a static charge at the point of contact. If there is no means of isolating this charge to the portion of the shaft where the sliding is occurring, it can pass through to the motor shaft and thence through the motor bearings.

Lastly – the frequency of the variable frequency drive harmonics in the output waveform is significantly higher than line frequency. This requires specific accommodations for grounding as traditional methods are insufficient due to the attenuation caused by the relatively high resistance ground path.